Methods and devices for real-time monitoring of tunable filters

ABSTRACT

Methods and devices suitable for monitoring the frequency of microwave tunable filters in real time. The frequency readout relies on the natural response of such a filter when excited by a pulse. Methods of measuring an operating frequency of a pole in a tunable filter include measuring a number of cycles in a natural response in the filter when the filter is excited by an electric current pulse, and determining a resonance frequency based on the number of cycles measured in the natural response. Such a method can provide the operating frequency information in a binary digital format, making it relatively easy to read and process. A measuring resonator may be mounted to the filter resonator and connected by a common actuator.

CROSS REFERENCE TO RELATED APPLICATIONS

The present patent application is a divisional of U.S. Utilityapplication Ser. No. 15/057,072, filed Feb. 29, 2016, which is relatedto and claims the priority benefit of U.S. Provisional PatentApplication Ser. No. 62/121,548, filed Feb. 27, 2015, the contents ofwhich are hereby incorporated by reference in their entirety into thepresent disclosure.

STATEMENT REGARDING FEDERALLY FUNDED RESEARCH

This invention was made with government support under Contract No.HR0011-12-C-0096 awarded by the Defense Advanced Research ProjectsAgency. The Government has certain rights in the invention.

TECHNICAL FIELD

The present application relates to microwave tunable filters and, morespecifically, to methods of monitoring and tuning the frequency ofhigh-Q microwave tunable filters in real time.

BACKGROUND

Tunable filters are the essence of emerging reconfigurable radios andspectrum-aware systems. Their capabilities of switching bands, changingcommunication standards, and handling jammers, among others, make them avery attractive choice for radio frequency (RF) front ends. Yet, theflexibility of tunable filters comes at the cost of being potentiallyvulnerable to variations in terms of frequency drift caused by aging orenvironmental effects. Such frequency stability issues can be addressedwith high-Q cavity filters that are tunable using equipment such asnetwork analyzers or by monitoring other operating modes, e.g.,secondary mode, in the cavity of the filter. However, these tuningmethods can be costly and time-consuming processes. Accordingly, thereis a need for improvements in the field.

SUMMARY

The present invention provides methods and devices suitable formonitoring the frequency of microwave tunable filters in real time. Thefrequency readout relies on the natural response of such a filter whenexcited by a pulse.

According to various aspects, an evanescent-mode RF filter is disclosed,comprising an RF filter resonator having a first membrane enclosing afirst cavity, a monitoring resonator having a second membrane enclosinga second cavity, the monitoring resonator mounted opposing the filterresonator such that the first and second membranes are facing oneanother, a planar actuator mounted between the first and secondmembranes, and a power supply configured to apply a voltage bias signalto the actuator, the voltage bias signal causing the actuator toincrease or decrease the operating frequency of the filter resonator.The filter may further comprise a pulse injection circuit operativelyconnected to an input of the monitoring resonator, the pulse injectioncircuit configured to supply a pulse signal to the monitoring resonator.The filter may further comprise a readout circuit connected to an outputof the monitoring resonator, the readout circuit configured to determinea number of pulses from the output having a voltage greater than apredetermined threshold in a predetermined time period, the number ofpulses corresponding to a natural response frequency of the filterresonator in response to the pulse signal.

Methods of measuring an operating frequency of a pole in a tunablefilter include measuring a number of cycles in a natural response in thefilter when the filter is excited by an electric current pulse, anddetermining a resonance frequency based on the number of cycles measuredin the natural response. Such a method can provide the operatingfrequency information in a binary digital format, making it relativelyeasy to read and process.

BRIEF DESCRIPTION OF THE DRAWINGS

In the following description and drawings, identical reference numeralshave been used, where possible, to designate identical features that arecommon to the drawings.

FIG. 1 is a diagram showing a communication system according to variousaspects.

FIG. 2a is a perspective view diagram of an RF filter monitoring systemaccording to various aspects.

FIG. 2b is a cross-sectional side view of a monitoring device accordingto various aspects.

FIG. 2c is a plot showing the relationship between resonator gaps for afilter resonator and monitoring resonator according to various aspects.

FIG. 2d is a plot showing the relationship between filter resonator andmonitoring resonator frequencies when the gap changes according tovarious aspects.

FIG. 3 is a plot showing an injected pulse to a monitoring resonatoraccording to various aspects.

FIG. 4 is a schematic diagram of a filter monitoring system according tovarious aspects.

FIG. 5 is a schematic block diagram of a feedback control circuitaccording to various aspects.

FIG. 6 is a schematic block diagram of a power supply circuit accordingto various aspects.

The attached drawings are for purposes of illustration and are notnecessarily to scale.

DETAILED DESCRIPTION

Various aspects relate to electrostatic control of an ionic environmentin a droplet based platform for biological applications. The terms “I,”“we,” “our” and the like throughout this description do not refer to anyspecific individual or group of individuals.

Throughout this description, some aspects are described in terms thatwould ordinarily be implemented as software programs. Those skilled inthe art will readily recognize that the equivalent of such software canalso be constructed in hardware, firmware, or micro-code. Becausedata-manipulation algorithms and systems are well known, the presentdescription is directed in particular to algorithms and systems formingpart of, or cooperating more directly with, systems and methodsdescribed herein. Other aspects of such algorithms and systems, andhardware or software for producing and otherwise processing signals ordata involved therewith, not specifically shown or described herein, areselected from such systems, algorithms, components, and elements knownin the art. Given the systems and methods as described herein, softwarenot specifically shown, suggested, or described herein that is usefulfor implementation of any aspect is conventional and within the ordinaryskill in such arts.

FIG. 1 shows a communication system 100, having an antenna 102 connectedto an radio frequency (RF) cavity filter 104, a feedback control circuit106 connected to the filter 104, and a receiver 108 connected to thefilter 104. In certain embodiments, the filter 104 may comprise anevanescent-mode cavity filter. In operation, the antenna 102 receivesradio frequency signals and directs them to the filter 104. The controlcircuit 102 tunes the filter 104 to a desired frequency or frequencies110 and holds the filter 104 at that frequency, regardless of effectsfrom hysteresis or creep. The control circuit 106 may operate on polesof the filter 104 independently, without interfering with the receivedRF signal. In certain embodiments, the control circuit 102 is configuredto tune the filter 104 with a resolution of 33 MHz to 6 MHz (3.5-0.4%)in the frequency range of 0.9-1.45 GHz. In other embodiments, theresolution may be 20 MHz to 2 MHz (0.13-1.3%). The frequency range mayinclude RF signals in the 1 GHz to 5 GHz range. The frequency range mayalso include microwave signals in the 300 MHz to 300 GHz range.

FIG. 2(a) shows a diagram of an evanescent-mode cavity filter 200 havinga monitoring device 201 for monitoring each pole the filter 200(illustrated here as a two pole filter). A cross-sectional side view ofthe monitoring device 201 is shown in FIG. 2(b). As shown, themonitoring device 201 comprises a monitoring cavity resonator 204stacked on top of each filter resonator 202 of the filter 200 in anopposing fashion. The monitoring resonator 204 comprises rigid housing218, a post 220 and a membrane 214 which encloses a cavity 222. Thefilter resonator 202 comprises a rigid housing 228, a post 224 and amembrane 216 which encloses a cavity 226. The bottom side of themembrane 214 of monitoring resonator 204 is mounted to a top side of anactuator, such as piezoelectric disk 208 to which a voltage bias isapplied to tune the filter 200. The bottom side of the piezoelectricdisk 208 is mounted to a top side of the membrane 216 of the filterresonator 202. It shall be understood that separate monitoring devices201 may be used for the different poles (filter resonators) in thefilter 200.

The piezoelectric disk 208 is electrically isolated from the membrane216 by an insulating material 210, which in one embodiment is anelectrically insulating glue. The glue is applied in a thin layer,allowing mechanical attachment without having a large impact on thetuning range of the piezoelectric disk 208. The piezoelectric disk 208may also be electrically insulated from the membrane 214.

Since the resonant frequency of each resonator 222 and 226 is controlledby the gap g (see FIG. 2(b)) between the post and the membrane (as shownin FIG. 2(c)), and since the gaps of both cavities 222 and 226 arecontrolled by the same actuator (e.g., piezoelectric disk 208), theresonant frequency of the monitoring resonator 204 (f_(mon)) will changewhenever the resonant frequency of the bottom filter cavity 226(f_(RFcav)) changes. Hence, monitoring the frequency of one cavityreveals the frequency of the other. This technique is not susceptible tohysteresis, creep or temperature effects, since any changes in onecavity will be reflected in the other.

The relationship between the resonant frequency of a cavity and its gapis given by

$\begin{matrix}{f \approx \frac{1}{2\;\pi\sqrt{L\; C}} \approx {\frac{1}{2\;\pi}\sqrt{\frac{g}{\epsilon_{0}A\; L}}}} & (1)\end{matrix}$where L and C are the effective inductance and capacitance of thecavity, respectively, g is the gap between the post and the membrane,and A is the area of the top of the post. The approximation in equation(1) is due to the parallel plate approximation of the capacitor.

From equation (1), the relationship between the resonant frequency of acavity and the gap between the post membrane is monotonic and bijective(one-to-one correspondence). By transitivity, the relationship betweenthe two resonant frequencies of the filter resonator 202 and monitoringresonator 204 is also monotonic and bijective. This relationship isshown in FIG. 2(d).

The frequency of the monitoring cavity 222 may be equal to, greater, orsmaller than the frequency of the RF filter cavity 226. This is also dueto the monotonic relationship between the two frequencies. In addition,since the monitoring and the RF paths are separated, each cavity can beoptimized independently.

In order to excite the monitoring resonator 204, a pulse injectioncircuit 402 is provided as shown in FIG. 4. The pulse injection circuit402 generates a current pulse 403 to excite the monitoring resonator204. In one embodiment, the pulse is generated by applying a stepwaveform on one input of an XOR gate 406, and a delayed version of thatstep waveform to the other input of the XOR gate 406. In one embodiment,an RC circuit consisting of a series resistance (R_(DELAY)) and theinput capacitance of the XOR gate 406 create the delay. The output ofthe XOR gate then drives a transistor 412 (shown here as an npntransistor, although other types may be used) through a current-limitingresistor 414 (R_(LIM)). The transistor 412 generates the current-pulse403 at the input of the monitoring resonator 204. An examplecurrent-pulse 403 output of the pulse injection circuit 402 is shown inFIG. 3.

The frequency of the monitoring resonator 204 can be detected from thenatural response of the filter. Therefore, the natural response shouldbe analyzed. In order to study the response of the cavity to a pulse,the cavity needs to be modeled. The monitoring cavity resonator 204 canbe modeled as a parallel RLC circuit 416, as shown in FIG. 4. When anRLC circuit is excited by a short current pulse (such as pulse 403), thenatural voltage response is a damped sinusoid. The voltage across ahigh-Q parallel RLC circuit under natural response can be approximatedas

$\begin{matrix}{V_{R\; L\; C} = {V_{0}e^{\frac{- t}{2\; R\; C}}{\sin\left( {{2\;\pi\; f_{0}t} + \theta} \right)}}} & (2)\end{matrix}$where V₀ is a constant, t is time, R and C are the resistance andcapacitance, f₀ is the natural frequency expressed in equation (1), andθ is the phase. This has been verified by simulating an RLC modelcircuit (using SPICE) when excited by the measured pulse from FIG. 3.

The current pulse from the pulse injection circuit 402 will typicallyexhibit jitter. Jitter can be caused by several mechanisms such asrandom additive noise. Additive noise can cause the logic gate (XOR 406)to trigger before or after the signal reaches the threshold, randomly.This causes different output pulse widths, which can change the responseof the circuit. As a result, the monitoring resonator 204 should bedesigned such that the response is not significantly affected by jitter.The frequency of the monitoring resonator 204 is chosen such that theresponse is not largely affected by the jitter in the pulse, whichbecomes prominent at frequencies close to the inverse of the pulsewidth. On the other hand, the frequency of the monitoring resonator 204cannot be too low because filter fabrication becomes problematic at lowfrequencies due to size requirements. As a result, the frequency of themonitoring resonator 204 should preferably be chosen between those twolimits. If the aforementioned limitations on the monitoring resonator204 define a range smaller than the tuning range (limited by thepiezoelectric actuator 208), the upper limit can be further moved tohigher frequencies by using a pulse injection circuit 402 that canprovide a smaller pulse width (T_(pulse)).

The natural frequency response of the monitoring resonator 204 is neededto determine the frequency of the filter resonator 202. When tuning thefilter 200, the capacitance C changes, which, in turn, changes thenatural response waveform in equation (2). This change can be detectedby counting the number of cycles above a voltage threshold in the dampedresponse, as shown in FIG. 4. This can be expressed analytically asN=f _(mon) t ₀  (3)where N is the number of cycles above the threshold, t₀ is the time ittakes for the signal to go below the threshold, and f_(mon) is thenatural resonant frequency of the monitoring resonator 204. Given thatthe sinusoidal component in equation (2) has a unity maximum, t₀ can befound by solving

$\begin{matrix}{{V_{0}e^{\frac{- t_{0}}{2\; R\; C}}} = V_{T}} & (4)\end{matrix}$where V_(T) is the threshold voltage. From equation (4), t₀ can be foundto be

$\begin{matrix}{t_{0} = {2\; R\; C\;{\ln\left( \frac{V_{0}}{V_{T}} \right)}}} & (5)\end{matrix}$

From equations (1), (3) and (5), the relationship between the number ofcycles and natural resonant frequency of the monitoring resonator 204 isgiven by

$\begin{matrix}{N = {\frac{2\; R}{\left( {2\;\pi} \right)^{2}f\;{mon}^{L}}{\ln\left( \frac{V_{0}}{V_{T}} \right)}}} & (6)\end{matrix}$

From equation (6), it can be seen that, in the natural response of acavity, the number of cycles that are above a voltage threshold (V_(T))is inversely proportional to the resonant frequency. This relationshipis also monotonic and bijective, which allows it to be used formonitoring.

Since the number of cycles N is inversely proportional to the monitoringresonator 204 frequency (Nα1/f_(mon)), and since the monitoringresonator 204 frequency is inversely proportional to the filterresonator 202 (f_(mon)α1/f_(RF cav)), the number of cycles N is directlyproportional to the filter resonator 202 frequency (Nαf_(RF cav)).

As shown in FIG. 4, a readout circuit 430 outputs the number of cyclesabove a voltage threshold in the signal output from the monitoringresonator 204. In one embodiment, the readout circuit 430 comprises alimiting amplifier 432 as the input stage. The limiting amplifier 432outputs a signal with a constant amplitude as long as the input islarger than the set threshold. The output of the limiting amplifier 432drives a high-speed ripple counter 434 to count the number of cycles.

When a pulse 403 is injected into the monitoring resonator 204, thecounter 434 provides the number of cycles observed in the dampedresponse. As discussed herein, the number of pulses can identify theresonant frequency of the monitoring resonator 204. As a result, thefrequency of the filter resonator 202 is determined as well.

In certain embodiments, the monitoring readout circuit 430 outputs thenumber of pulses output from the monitoring resonator 204 in digitalform, easing the integration of the readout in a control system.

FIG. 5 shows one embodiment of the feedback control circuit 106 whichprovides tuning of each pole in the filter 200 to a desired frequency,and to maintain that tuning regardless of memory effects such ashysteresis or creep. To accomplish this, the control circuit 106 changesthe power supply that generates the bias voltage of the piezoelectricdisk 208 based on the frequency reading from the readout circuit 430.

As shown in FIG. 5, the control circuit 106 takes two inputs, thereadout (N) from the readout circuit 430 and a digital number (D_(IN))representing the desired operating frequency. The input received fromthe readout circuit 430 is first averaged (using averaging unit 502) tosuppress any noise in the monitoring reading. Experiments show thataveraging over 32 k samples seems sufficient for the data to be stableand flicker free. Also, the data input (D_(IN)) is latched (using latch504) and sent to a look-up-table (LUT unit 506) to generate an initiallyestimated control signal to the power supply (PS_(CT RL) est.). Thisspeeds up the process of generating the correct control signal to thepower supply. The averaged readout data are then compared with D_(IN)using a binary magnitude comparator 508. If they are not equal, asdesired, a counter 510 generates an error signal (positive or negative)which will be added to the estimated signal. This will change controlsignal (PS_(CTRL)) of the power supply 512 (which is connected to thepiezoelectric disk 208 to supply the voltage bias to the disk 208). Theerror signal will keep increasing (or decreasing if negative) until theaveraged readout data is equal to the desired input data (D_(IN)). Atthat point, the power supply control signal PS_(CTRL) has adjusted thepower supply 512 to generate the piezoelectric bias signal V_(Bias) thatwould correspond to the desired operating frequency of the filter 200.If the frequency of the filter resonator 202 is changed due to creep orany other environmental perturbations, it will change the readout signal(N). This will cause the control circuit 106 to change the error signaluntil the operating frequency of the filter 200 is correctedautomatically.

In certain embodiments, the control circuit 106 is fully digital cantherefore be implemented in a microcontroller or a field programmablegate array (FPGA) platform.

In certain embodiments, to ease the integration of the system, the powersupply 512 may be controlled digitally and should be capable ofgenerating high voltage bias for the piezoelectric disk 208. FIG. 6shows one embodiment of the power supply 512 which comprises aDigital-to-Analog converter (DAC) 602 and amplifier 604. The DAC 602receives the power supply control signal (PS_(CTRL)) and converts thesignal to an analog low-voltage replica of the desired voltage. Theoutput of DAC 602 is directed to the high voltage amplifier 604. Theoutput of the amplifier 604 is then directed to the piezoelectric disk208 as shown in FIG. 6.

Steps of various methods described herein can be performed in any orderexcept when otherwise specified, or when data from an earlier step isused in a later step. Exemplary method(s) described herein are notlimited to being carried out by components particularly identified indiscussions of those methods.

According to various aspects, technical effects can include thecapability of measuring an operating frequency of a pole of a filter inreal time with relatively low cost devices. In preferred embodiments,the frequency response of each pole in a filter can be measured usingsimple circuitry using off-the-shelve electronics that can be embeddedin a system with reduced power consumption overhead, resulting in arelatively inexpensive solution in comparison to conventional techniquestuned with lab equipment. Also, these methods preferably provide thefrequency information in a digital format, and without affecting themain cavity operation.

Various aspects described herein may be embodied as systems or methods.Accordingly, various aspects herein may take the form of an entirelyhardware aspect, an entirely software aspect (including firmware,resident software, micro-code, etc.) run by one or more computerprocessors connected to electronic memory, or an aspect combiningsoftware and hardware aspects These aspects can all generally bereferred to herein as a “service,” “circuit,” “circuitry,” “module,” or“system.”

Furthermore, various aspects herein may be embodied as computer programproducts including computer readable program code (“program code”)stored on a computer readable medium, e.g., a tangible non-transitorycomputer storage medium or a communication medium. A computer storagemedium can include tangible storage units such as volatile memory,nonvolatile memory, or other persistent or auxiliary computer storagemedia, removable and non-removable computer storage media implemented inany method or technology for storage of information such ascomputer-readable instructions, data structures, program modules, orother data. A computer storage medium can be manufactured as isconventional for such articles, e.g., by pressing a CD-ROM orelectronically writing data into a Flash memory. In contrast to computerstorage media, communication media may embody computer-readableinstructions, data structures, program modules, or other data in amodulated data signal, such as a carrier wave or other transmissionmechanism. As defined herein, “computer storage media” do not includecommunication media. That is, computer storage media do not includecommunications media consisting solely of a modulated data signal, acarrier wave, or a propagated signal, per se.

The invention is inclusive of combinations of the aspects describedherein. References to “a particular aspect” (or “embodiment” or“version”) and the like refer to features that are present in at leastone aspect of the invention. Separate references to “an aspect” (or“embodiment”) or “particular aspects” or the like do not necessarilyrefer to the same aspect or aspects; however, such aspects are notmutually exclusive, unless otherwise explicitly noted. The use ofsingular or plural in referring to “method” or “methods” and the like isnot limiting. The word “or” is used in this disclosure in anon-exclusive sense, unless otherwise explicitly noted.

The invention has been described in detail with particular reference tocertain preferred aspects thereof, but it will be understood thatvariations, combinations, and modifications can be effected within thespirit and scope of the invention.

The invention claimed is:
 1. A method of measuring an operatingfrequency of a pole in an RF filter, the method comprising: providing amonitoring resonator opposedly mounted to a filter resonator of the RFfilter, the monitoring resonator having a monitoring resonator membrane,the filter resonator having a filter resonator membrane, such that themonitoring resonator membrane and filter resonator membrane areconnected by a common planar actuator; injecting a pulse into themonitoring resonator using a pulse injection circuit operativelyconnected to an input of the monitoring resonator, the pulse injectioncircuit configured to supply a pulse signal to the monitoring resonator;and measuring a number of output pulses having a voltage greater than apredetermined threshold in a predetermined time period from an output ofthe monitoring resonator using a readout circuit connected to the outputof the monitoring resonator to determine a number of cycles in a naturalresponse of the filter, wherein the readout circuit outputs a value in abinary digital format, the value corresponding to said number of pulses;determining a resonance frequency based on the number of cycles measuredin the natural response.
 2. The method according to claim 1, wherein theoperating frequency is provided in a binary digital format.
 3. Themethod of claim 1, further comprising: directing the output of thereadout circuit to a feedback control circuit, the feedback controlcircuit having an input configured to receive output from the monitoringresonator and an output operatively connected to a power supplyconnected to the filter resonator, the feedback control circuitconfigured to tune the filter to a desired operating frequency.
 4. Themethod of claim 3, wherein the planar actuator comprises a piezoelectricelement.
 5. The method of claim 4, wherein the piezoelectric element iselectrically isolated from the filter resonator membrane by a firstelectrically insulating material.
 6. The method of claim 5, wherein thepiezoelectric element is electrically isolated from the monitoringresonator membrane by a second electrically insulating material.
 7. Themethod of claim 6, wherein the first electrically insulating materialand the second electrically insulating material are the same.
 8. Themethod of claim 4, wherein the piezoelectric element comprises a disc,and wherein the filter resonator and the monitoring resonator arecircular.
 9. The method of claim 3, wherein the filter is a band-stopfilter.
 10. The method of claim 1, wherein the filter comprises aplurality of filter resonators and monitoring resonators, each of thefilter resonators connected to a corresponding monitoring resonator.